Solid state power circuits employing
new autoimpulse commutation

ABSTRACT

THIS INVENTION COMPRISES A FAMILY OF IMPROVED POWER CIRCUITS USING A PAIR OF TURN-ON, NONGATE TURN-OFF CONTROLLED CONDUCTING DEVICES INTERCONNECTED IN SERIES CIRCUIT RELATIONSHIP WITH A TAPPED FIRST LINEAR INDUCTOR ACROSS A PAIR OF POWER SUPPLY TERMINALS THAT ARE ADAPTED TO BE CONNECTED ACROSS A SOURCE OF ELECTRIC POTENTIAL. A LOAD CIRCUIT IS SEPARATELY CONNECTED IN SERIES CIRCUIT RELATIONSHIP WITH THE FIRST OF SAID PAIR OF DEVICES WITH THE LOAD CIRCUIT BEING CONNECTED THROUGH A DIRECT CURRENT PATH BETWEEN THE TAP POINT ON THE FIRST LINEAR INDUCTOR AND ONE OF THE POWER SUPPLY TERMINALS. THE CIRCUIT ARE FURTHER COMPRISED BY COMMUNICATION CIRCUIT MEANS FORMED BY AT LEAST ONE COMMUTATING CAPACITANCE AND SERIES CONNECTED SECOND LINEAR INDUCTOR WITH THE SERIES CIRCUIT THUS COMPRISED BEING CONNECTED BETWEEN THE TAP POINT ON THE FIRST LINEAR INDUCTOR AND TO THE REMAINING TERMINAL (NOT CONNECTED TO THE FIRST INDUCTOR) OF ONE OF THE CONDUCTING DEVICES. THE SERIES CIRCUIT COMPRISED BY THE COMMUTATING CAPACITOR AND SECOND INDUCTOR IS TURNED TO SERIES RESONANCE AT A COMMUNICATING FREQUENCY HAVING A PERIOD WHICH IS SUBSTANTIALLY SHORTER   THAN THE LOAD CURRENT CONDUCTING PERIODS OF THE POWER CIRCUIT. WITH THIS ARRANGEMENT, ONE OF THE PAIR OF CONTROLLED CONDUCTING DEVICES IS ADAPTED TO BE INTERMITTENTLY RENDERED CONDUCTIVE FOR DISCHARGING THE COMMUTATING CAPACITANCE THROUGH A PORTION OF THE FIRST INDUCTOR AND THEREBY TERMINATING THE CONDUCTION OF THE OTHER OF THE PAIR OF CONDUCTING DEVICES. IN PREFERRED EMBODIMENTS BIDIRECTIONAL CONDUCTING DEVICES ARE EMPLOYED, AND WHERE SUCH BIDIRECTIONAL DEVICES ARE EMPLOYED THE CIRCUITS MAY ALSO PROVIDE OPERATION IN TWO DIFFERENT MODES TO ALLOW FOR POWER SUPPLY TO THE LOAD AND FOR PUMP BACK TO POWER FROM THE LOAD TO THE POWER SOURCE, AS WELL AS INVERSION FROM DIRECT CURRENT TO ALTERNATING CURRENT. BRIDGE TYPE POWER CIRCUITS COMPRISING THE SAME BASIC CIRCUIT CONFIGURATIONS ARE ALSO PROVIDED.

June 1, 1971 R, g, MORGAN AL R 7,128

SOLID STATE POWER CIRCUITS EMPLOYING NEW AUTOIMPULSE COMMUTATIONOriginll File dlaroh 2s, 1.964 15 Sheets-Sheet 1 M K IMF/46: i

s/ cwaio [/7 vent ans Paymona i/Vor an, fiur'n/ce flfiea 0rd,

The/r Attorney.

R. E. MORGAN ETAL Re. 21,128 SOLID STATE POWER CIRCUITS EMPLOYING June1, 1 971 NEW AUTOIMPULSE COMMUTATION Original Filed March 26,, 1964 1.5Sheets-Sheet [)7 1 6)? 6 0215. Raymond ZT/Vo Barn/cc flied o d,

. 777w)" Attorney.

.June 1, 1971' R. a. MORGAN ETAL Re. 27,128

SOLID STATE POWER CIRCUITS EMPLOYING NEW AU'I'OIMPULSE COMMU'I'ATION 15Sheets-Sheet 4 Original Filed march 26. 1964 [r7 ve r7 Cons: PaymandMeagan, Z3 ur'n/ce .0. Be aforaf The/r At borney.

June 1, 1971 MORGAN EIAL Re. 27,128

SOLID STATE POWER CIRCUITS EMPLOYING NEW AUTOIMPULSE COMMUTATION IOriginal Filed March 26, 1964 15 Sheets-Sheet b [n ven 25 ans.- Paymondf Morgan, Burn/ca fi. B eafor-d,

June 1, 1971 R MORGAN ETAL Re. 27,128

SOLID STATE POWER CIRCUITS EMPLOYING NEW AU'I'OIMPULSE COMMUTATIONOriginal Filed larch 26, 1964 15 Sheets-Sheet 7 S if i V i: 65 4 ,R

l l [S L ..Jj'\ I 7 c 9c 6 q 6 .53: z/ z I 1 6 0% 74 22 E z J F7 ./6'[f7 veflcorzsx 5 Raymond Morgan, 5 urn/cc fl Bea ford The/f" Az: Carney.

June 1, 1971 MORGAN ETAL Re. 27,128

SOLID STATE POWER CIRCUITS EMPLOYING NEW AUTOIMPULSE COMMUTATIONOriginal Filed March 26, 1964 15 Sheets-Sheet 8 [n vent or's: Fag mafiaf. Mor an, fiz/r'n/ce flBe foraf 777w)" A a: t orney June 1, 1971"MORGAN ETAL Re. 27,128

SOLID STATE POWER CIRCUITS EMPLOYING NEW AUTOIMPULSE COMMUTATIONOriginal Filed larch 26, 1964 15 Sheets-Sheet 9 W W M mzfli/m e d mmm m3 w; R5

June 1, 1971 5, MQRGAN EIAL Re. 21,128

. SOLID STATE POWER CIRCUITS EMPLOYING NEW AUTOIMPULSE COMMUTATIONOriginal Filed llaroh 26, 1964 us Sheets-Sheet 10,

5 away/r 92 c I I a A 0 4fi [3 a/Pca/r [/7 vent ans.-

Ragmano Morgan.

Born/Ce 256070112, by )Z/ 4 M 7770)" Attorney.

June 1, 1971 MORGAN EI'AL Re. 27,128

SOLID STATE POWER CIRCUITS EMPLOYING NEW AUTOIMPULSE COMMUTATIONOriginal Filed March 26, 1964 15 Sheets-Sheet 1! 1 0A 0 C/IfCW/f [)7vent 011s: Fag/m QfldfMOflQ an,

by 94! 4. M

The/k Attorney.

June 1, 1971 MORGAN ETAL Re. 27,128

SOLID STATE FOWER CIRCUITS EMPLOYING NEW AU'I'OIMPULSE COMMUTATIONOriginal Filed larch 26, 1964 15 Sheets-Sheet 18 53%, v Q SWQN w M ZMW Mi c .5 76 nf A W r "ac u} 6 a0 7 PB June 1, 1971 R, MORGAN EI'AL Re.21,128

$01.11) STATE POWER CIRCUITS EMPLOYING NEW AUTOIMPULSE COMMU'IATIONOriginal Filed larch 26, 1964 15 Sheets-Sheet 1 x m fi i MW sa 0 #nf slym W 8 5g xvw sh Q m g Q \P NQEN N Q $4 7776/)" A orpey H United StatesPatent 27,128 SOLID STATE POWER CIRCUITS EMPLOYING NEW AUTOIMPULSECOMMUTATION Raymond E. Morgan, deceased, late of Schenectady, N.Y.,

by Agnes T. Morgan, administratrix, Schenectady, and

Burm'ce D. Bedford, Scotia, N.Y., assignors to General Electric CompanyOriginal No. 3,376,492, dated Apr. 2, 1968, Ser. No.

354,888, Mar. 26, 1964. Application for reissue Apr.

1, 1969, Ser. No. 812,473

Int. Cl. H02m 7/48 US. Cl. 32143 71 Claims Matter enclosed in heavybrackets appears in the original patent but forms no part of thisreissue specification; matter printed in italics indicates the additionsmade by reissue.

ABSTRACT OF THE DISCLOSURE This invention comprises a family of improvedpower circuits using a pair of turn-on, nongate turn-off controlledconducting devices interconnected in series circuit relationship with atapped first linear inductor across a pair of power supply terminalsthat are adapted to be connected across a source of electric potential.A load circuit is separately connected in series circuit relationshipwith the first of said pair of devices with the load circuit beingconnected through a direct current path between the tap point on thefirst linear inductor and one of the power supply terminals. Thecircuits are further comprised by commutation circuit means formed by atleast one commutating capacitance and series connected second linearinductor with the series circuit thus comprised being connected betweenthe tap point on the first linear inductor and to the remaining terminal(not connected to the first inductor) of one of the conducting devices.The series circuit comprised by the commutating capacitor and secondinductor is tuned to series resonance at a commutating frequency havinga period which is substantially shorter than the load current conductingperiods of the power circuit. With this arrangement, one of the pair ofcontrolled conducting devices is adapted to be intermittently renderedconductive for discharging the commutating capacitance through a portionof the first inductor and thereby terminating the conduction of theother of the pair of conducting devices. In preferred embodimentsbidirectional conducting devices are employed, and where suchbidirectional devices are employed the circuits may also provideoperation in two different modes to allow for power supply to the loadand for pump back of power from the load to the power source, as well asinversion from direct current to alternating current. Bridge type powercircuits comprising the same basic circuit configurations are alsoprovided.

Our invention relates to a family of new and improved power circuitsemploying new controlled turn-on conducting devices and a new andimproved turn-off or commutation means therefor.

More particularly, our invention relates to a family of power circuitsemploying turn-on, nongate turn-oft solid state semiconductor controlleddevices for power switching purposes and is especially useful intime-ratio control of direct current electric power or for inversion ofdirect current electric power to alternating current electric power.Time-ratio control of direct current electric power refers to theinterruption or chopping-up of a direct current electric potential bycontrolling the on time of a turn-on, turn-off power switching deviceconnected in circuit relationship with a load and the direct currentelectric potential. Inversion of direct current electric Re. 27,128 Reissuecl June 1, 1971 power to alternating current electric power refersto the switching of a load across alternate output terminals of a directcurrent electric supply by appropriately switching turn-on, turn-offpower switching devices connecting the load in circuit relationship withthe direct current electric supply.

In recent years, the turn-on, turn-off power switching devices employedin the above described types of power circuits for the most part haveemployed a solid state semiconductor device known as a siliconcontrolled rectifier (SCR). The SCR is a four-layer PNPN junction devicehaving a gating electrode which is capable of turnlng on current flowthrough the device with only a relatively small gating signal. Theconventional SCR, however, is a nongate turn-off device in that onceconduction through the device is initiated, the gate thereafter losescontrol over conduction through the device until it has been switchedoff by suitable external means. Such external means are generallyreferred to as commutation circuits and usually effect commutation orturning off of the SCR by reversal of the potential across the SCR. Inaddition to the SCR, recent advances in the semiconductor art have madeavailable to industry new solid state semiconductor devices which arecontrolled turn-on, nongate turn-off conducting devices, but which arebidirectional conducting devices. A bidirectional conducting device is adevice capable of conducting electric current in either directionthrough the device. The first of these devices, referred to as a triac,"is a gate controlled turn-on NPNPN junction device which, similar to theSCR, is a nongate turn-off device that must be turned off by externalcommutation circuit means. While the preferred form of a triac is afive-layer gate controlled device, it should be noted that four-layerPNPN and NPNP junction gate controlled triac devices are practical, aswell as other variations, but the triac characteristics mentioned aboveare common to all. The second newly available power device, referred toas a power diac is a twoterminal, five-layer NPNPN junction devicewhich, like the triac, has bidirectional conducting characteristics. Incontrast to the SCR and triac, however, the diac is not a gate turn-ondevice, but must be turned on by the application of a relatively steepvoltage pulse (high dv/dt applied across its terminals. It should benoted that the SCR and triac may also be fired by the same high dv/dttechnique. However, the diac is similar to the SOR and triac in that ittoo must be turned off by external circuit commutation means. Ourinvention provides new and improved power circuits employing solid statesemiconductor devices of the above general type as well as a new andimproved commutation scheme for use with such devices. It should beexpressly noted in this regard that the term nongate turn-off device asemployed hereinafter and in the claims, is intended to include not onlythe specific devices discussed above but also includes so called gateassisted turn-off devices (also referred to as a GTOSCR) which requireexternal commutation circuit means to assure complete turn-off, althoughthe device is capable of achieving some degree of turn-off by theapplication of a reverse polarity, turn-off signal to its control gate.Additionally, it should be noted that the generic term bidirectionalconducting device as employed hereinafter and in the claims, is intendedto cover not only the single triac and diac bidirectional conductingdevices described briefly above, but also is intended to cover suchknown arrangements as reverse polarity, parallel connected SCRs as wellas a single SCR and reverse polarity, parallel connected diode, etc.These devices are to be distinguished from controlled conductivitybidirectional conducting devices" (such as triacs, diacs, reversepolarity, parallel connected SCRs) wherein conduction 3 tough the devicein each directionis controlled. Power 'cuits employing bidirectionalconducting devices have en disclosed in the published literature as wellas heretfter along with such circuits employing conductivity ntrolledbidirectional conducting devices.

It is, therefore, a primary object of our invention to ovide an entirefamily of new and improved power cirits employing controlled turn-onnongate turn-ofi" con- .cting devices. Another object of our inventionis to provide a new .d improved commutation scheme for power circuitsiploying controlled turn-on, nongate turn-off conductg devices whichallows for a reduction in the size of mponents employed in the circuitfor a given power ting and, hence, is economical to manufacture. Afurther object of our invention is to provide a new d improvedcommutation scheme which is economical d efficient in operation andwhich provides reliable mmutation that is independent of load from noload full load operating conditions. In practicing our invention, newand improved power cuits are provided using controlled turn-on, nongatern-otf solid state semiconductor devices. These new and tproved powercircuits include in combination a pair of terconnected turn-on, nongateturn-off controlled conlcting devices in series circuit relationshipacross a pair power supply terminals that, in turn, are adapted to bennected across a source of electric potential. The pair controlledconducting devices are interconnected by eans of a tapped first linearinductance. A first of the .ir of controlled conducting devices is alsoconnected series circuit relationship with a load circuit includingfilter inductance wherein the load circuit is connected tween the tappoint of the first linear inductance and 1e of the power supplyterminals. Turn-on gating and ing circuit means are provided forcontrolling the rn-on of the controlled conducting devices, andcomutation circuit means are provided for commutating off e devices atdesired intervals. The commutation cir- .it means comprises the tappedfirst linear inductance 1d a pair of series connected second linearinductances 1d commutating capacitors wherein a first of the pair ofries connected inductances and capacitors is connected tween the tappoint of the first inductance and a first the power supply terminals andthe second of the pair series connected inductances and capacitors iscon- :cted between the same tap point and the second power pplyterminal. Each of the pair of series connected seeid inductances andcapacitors is tuned to series resonance a substantially higher frequencythan the power cirit operating frequency. The features of our inventionwhich we desire to protect :rein are pointed out with particularity inthe appended aims. The invention itself, however, both as to itsganization and method of operation, together with furer objects andadvantages thereof, may thus be under- )od by reference to the followingdescription taken in nnection with the accompanying drawings whereinlike Its in each of the drawings are identified by the same .aracterreference and wherein: FIGURE 1 is a detailed circuit diagram of a newand lproved time-ratio control power circuit employing a :w and improvedcommutation means in accordance th our invention; FIGURE 2 is anequivalent circuit representation ilstrating the time-ratio controlprinciple together with series of curves depicting the form of variablevoltage rect current electric energy derived from time-ratio con- )1power circuits; FIGURE 3 is an equivalent circuit diagram of a timetiocontrol circuit and associated characteristic curves ustrating theeffect of a coasting rectifier and filter inlctance added to theequivalent circuit of FIGURE 2; FIGURE 4 is a detailed circuit diagramof a suitable .ting on circuit for use with the time-ratio control cir-.it shown in FIGURE 1;

FIGURE 5 is a detailed circuit diagram of a modification of the gatingcircuit shown in FIGURE 4 to provide independent control over thecommutation operation as well as independent control of the turn on ofthe load current;

FIGURE 6 is a detailed circuit diagram of a modification of the circuitshown in FIGURE 1 and employs triacs in place of the silicon controlledrectifiers and diodes;

FIGURE 7 is a detailed circuit diagram of the circuit shown in FIGURE 6including the details of the triac gate firing circuits;

FIGURE 8 is a detailed circuit diagram of a new and improved time-ratiocontrol circuit employing dv/dt fired SCRs and a new and improvedcommutation scheme comprising a part of our invention;

FIGURE 9 is a detailed circuit diagram of a modification of the circuitshown in FIGURE 8 and employs bidirectional conducting diacs in place ofthe dv/dt fired SCRs and, in addition, illustrates a different form ofcapacitor isolation between the two firing circuits;

FIGURE 10 is a modification of the circuit shown in FIGURE 8 and uses abidirectional conducting diac in place of one of the dv/dt fired SCRsand, in addition, illustrates another different form of capacitorisolation between the firing circuits;

FIGURE 11 is a detailed diagram of a new and improved time-ratio controlpower circuit incorporating many of the features of the circuit shown inFIGURE 9 and, in addition, illustrates a different form of firingcircuit means for turning on a diac or a dv/dt fired SCR;

FIGURE 12 is a detailed circuit diagram of still a different form offiring circuit means for turning on a diac which uses common circuitelements to turn on the diac to conduct current in either one of twoopposite directions;

FIGURE 13 is a modification of the circuit shown in FIGURE 12 whichprovides independent control of the turn-on of the bidirectionalconducting diac in either direction;

FIGURE 14 is a modification of the time-ratio control power circuitshown in FIGURE 11 with the exception that a bidirectional conductingtriac is substituted for one of the diacs of FIGURE 11;

FIGURE 15 is a modification of the time-ratio control power circuitshown in FIGURE 11 with the exception that a conventional gate fired SCRis substituted for one of the diacs of FIGURE 11;

FIGURE 16 is a modification of the time-ratio control power circuitshown in FIGURE 7 with the exception that a bidirectional conductingdiac is substituted for one of the triacs of FIGURE 7;

FIGURE 17 is a modification of the time-ratio control power circuitshown in FIGURE 7 with the exception that a dv/dt fired SCR and feedbackdiode are substituted for one of the triacs of FIGURE 7 and, inaddition, illustrates a different form of firing circuit for diacs anddv/dt fired SCRs;

FIGURE 18 is a modification of the time'ratio control power circuitshown in FIGURE 11 with the exception that a dv/dt fired SCR andfeedback diode are substituted for one of the diacs of FIGURE 11;

FIGURE 19 is a generalized circuit diagram of the time-ratio controlpower circuit constructed in accordance with our invention;

FIGURE 20 is a detailed circuit diagram of a new and improved powercircuit employing dv/dt SCRs and our new and improved commutation schemewherein the power circuit is operable either as a time-ratio controlpower circuit or single-phase inverter circuit depending upon theparticular sequence of firing the dv/dt SCRs;

FIGURE 21 is a detailed circuit diagram of a new and improvedsingle-phase inverter circuit employing the new and improved commutationscheme of our invention and using a triac and gate controlled SCR;

FIGURE 22 is a modification of the power circuit shown in FIGURE withthe exception that a triac is substituted for the first dv/dtSCR-feedback diode combination, a diac is substituted for the seconddv/dt SOR- coasting diode combination, and our new and improvedcommutation circuit is rearranged;

FIGURE 23 is a modification of the inverter circuit shown in FIGURE 21with the exception that a dv/dt SCR is substituted for the conventionalgate controlled SCR of FIGURE 21 and our commutation circuit is furtherrearranged;

FIGURE 24 is a detailed circuit diagram of a second form of the new andimproved single-phase inverter circuit constructed in accordance withour invention;

FIGURE is a detailed circuit diagram of a threephase inverter employingas its basic building block a circuit similar to the single-phaseinverter of FIGURE 22;

FIGURE 26 is a detailed circuit diagram of a singlephase, full-wavebridge inverter circuit employing as its basic building block a circuitsimilar to the single-phase inverter of FIGURE 23;

FIGURE 27 is a detailed circuit diagram of a modified version of thefull-wave bridge inverter circuit of FIG- URE 26;

FIGURE 28 is a detailed circuit diagram of still a third form ofsingle-phase, full-wave bridge inverter circuits employing as its basicbuilding block a circuit similar to the single-phase inverter of FIGURE21;

FIGURE 29' is a detailed circuit diagram of a new and improvedsingle-phase, full-wave bridge inverter circuit employing as its basicbuilding block the circuit shown in FIGURE 7;

FIGURE is a detailed circuit diagram of a new and improved single phase,full-wave bridge inverter circuit employing as its basic building blockthe circuit shown in FIGURE 11; and

FIGURE 31 is a modification of the power circuit shown in FIGURE 22.

A new and improved time-ratio control power circuit illustrated inFIGURE 1 of the drawings is comprised by a first gate turn-on, nongateturn-off solid state silicon controlled rectifier device, SCR 11, and aload 12, effectively coupled in series circuit relationship across apair of power supply terminals 13 and 14 which, in turn, are adapted tobe connected across a source of electric potential. In the particularembodiments of the invention shown herein, the source of electricalpotential E is a direct current power supply having its positivepotential applied to terminal 13 and its negative potential applied toterminal 14. It should be noted that While the timeratio controlcircuits herein disclosed are drawn in connection with direct currentpower supplies, with very little modification these circuits could beused to remove or chop out any desired portion of a half-cycle ofapplied alternating current potential. A filter inductance 15 isconnected in series circuit relationship intermediate SCR 11 and load 12and a second gate turn-on, nongate turnotf solid state SCR device 16 anda coasting diode 17 are connected in parallel circuit relationship withthe filter inductance 15 and load 12.

Commutation circuit means are provided for terminating the conduction(turning 013?) of SCR 11 and comprise tightly coupled tapped firstlinear inductance winding 18, which interconnects SCR 11 and SCR 16, anda pair of series connected second linear inductances and commutatingcapacitors. The first of the pair of series connected inductances andcapacitors, comprising inductance 19 and capacitor 20, is connectedbetween the tap point of inductance 18 and power supply terminal 13. Thesecond of the pair of series connected inductances and capacitors,comprising inductance 21 and capacitor 22 (shown in dotted line form),is connected between the tap point and the negative power supplyterminal 14. Each of the pair of series connected second inductances andcapacitors is tuned to series resonance at a frequency which issubstantially higher than the power circuit operating frequency. Theseries circuit comprising linear inductance 21 and commutating capacitor22 is shown in dotted line form since such circuit would not be requiredin the event that the direct current power source supplies an infiniteor stiff bus, that is, maintains a constant output voltage. In the moregeneral case, such output voltage is slightly variable and in such case,inductance 21 and capacitor 22 would be connected as shown. Properlyphased gating on signals are applied to the gating on electrodes of SCRs11 and 16 from a suitable gating signal control circuit such as thatshown in FIGURE 4 of the drawings for gating on the SCRs in properlytimed sequence as explained hereinafter. A second unidirectionalconducting device comprising feedback diode 23 may be directly connectedacross SCR 11 and is shown in dotted line form since this diode is usedonly when electric energy is being fed back to the power supply as alsoexplained hereinafter.

In operation, if it is assumed that initially SCR 11, which for purposesof explanation will be defined as the load current carrying SCR, and SCR16, which for this purpose will be described as the commutating SCR, areeach in their nonconducting or blocking state'but with coasting diode 11conduction due to a preceding cycle of operation, then capacitor 20 ischarged to the power supply voltage and the capacitor 22 has no chargethereon. The circuit remains in this condition until such time that agating on signal is applied to the gating on electrode of SCR 11. Uponthis occurrence, SCR 11 becomes conducting or turned on, current i issupplied thereto from the power supply, and the full power supplyvoltage E is essentially across inductance 18 with coasting diode 17conducting (due to a preceding cycle of operation). It will be assumed,for purposes of explanation, that winding 18 is center-tapped, althoughin the most general case the tap point need not be at the center. It,therefore, follows that the center tap of inductance 18 is at one-halfof the power supply voltage E This immediate rise of voltage at thecenter tap from 0 to /2 of the supply voltage causes capacitor 22 tobegin to charge and capacitor 20 to discharge. At steady stateconditions, the load current I flows in the series circuit comprisingSCR 11, the upper half of winding 18, filter inductance 15, and load 12.At such steady state conditions, the center tap of inductance 18, thedot end of capacitor 20, and the dot end of capacitor 22 are at fullline voltage. Load current carrying SCR 11 remains conducting for a timeperiod dependent upon the amount of current to be supplied to load 12and then is rendered nonconducting or commutated 01f in the manner of atime-ratio control power circuit.

The theory of operation of time-ratio power control is best illustratedin FIGURE 2 of the drawings wherein FIGURE 2(a) shows an on-off switch24 connected in series circuit relationship with a load resistor 25across a direct current power supply E With the arrangement of FIGURE2(a), there are two possible types of operation in order to supplyvariable amounts of power to the load resistor 25. In the first type ofoperation, switch 24 is left closed for fixed periods of time and thetime that switch 24 is left open can be varied. This type of operationis illustrated in curves 2(b), wherein curve 2(b)(1) illustrates acondition where switch 24 is left open for only a short period of timecompared to the time it is closed to provide an average voltage acrossload resistor 25 equal to about three-fourths of the supply voltage E ofthe direct current power supply. In FIGURE 2(b)(2) the condition isshown where the switch 24 is left open for a period of time equal tothat during which it is closed. Under this condition of operation, thevoltage across the load will equal approximately 50 percent of thesupply voltage E FIGURE 2(b)=(3) illustrates the condition where switch24 is left open for a period of time equal to three times that for whichthe switch is closed so that the load voltage appearing across e loadresistor 25 will be equal to about 25 percent the supply voltage E Itcan be appreciated that by rying the period of time during which switch24 is it open, the amount of direct current potential applied ross load25 is varied proportionally. In the second type of operation possiblewith time-ratio 'ntrol circuits, switch 24 is closed at fixed times, andthe he that the switch is left closed can be varied. This cond type ofoperation of the circuit shown in FIGURE a) is illustrated in FIGURE2(c) of the drawings aerein the amount of time that switch 24 is leftclosed varied. In FIGUR'E 2(c)(l), the condition where Iitch 24 is leftclosed for a much greater period of time an it is open, is illustratedto provide a load voltage of approximately 0.75E In FIGURE 2(c) (2), thene that switch 24 is left closed equals the time that it open to producea load voltage E that is equal to 9E In FIGURE 2(c)(3), the condition isillustrated here switch 24 is left closed for a period of time equalone-third of the time that switch 24 is left open to prode a loadvoltage equal to 0.25E It can be appreciated, erefore, that by varyingthe period of time that switch I is left closed, the amount of voltagesupplied across ad resistor 25 can be varied proportionally. In a simllfashion to that described with respect to switch 24, I varying theperiod of time that SCR 11 of the circuit own'in FIGURE 1 is either in aconducting or nonnducting condition, the power supplied to load 12 canvaried proportionally. It is a matter of adjustment of e phasing of thegating control signals supplied to the lntrol gates of SCR 11 and SCR 16which determines the nount of time that SCR 11 is either conducting ornonnducting. This of course, in turn, determines the power pplied toload 12 in the manner described with rela- )n to FIGURE 2. Whether theamount of time that SCR t is in its blocking condition is varied, orwhether the nount of time that SCR 11 is conducting is varied, to 'ovidesuch proportionally controlled power to load 12 ually depends upon theload in question. Insofar as the 'inciples of commutation to bedescribed hereinafter e concerned, it does not matter which type ofoperation employed. FIGURE 3 of the drawings better depicts the naturethe output signal or voltage E developed across load sistor 12 by thecircuit shown in FIGURE 1. In FIG- RE 3(a), SCR 11 is again depicted bythe on-oif switch I, and the voltage or current versus time curves forthe irious elements of this circuit are illustrated in FIGURE 1b).FIGURE 3 (b) (1) illustrates the voltage versus time iaracteristics ofthe potential e appearing across the asting diode 17. It is to be notedthat the potential is essentially a square wave potential whose periodis :termined by the timing of switch 24. For the period of no thatswitch 24 is left closed, a load current i flows rough filter inductance15, load 12, and back into the )wer supply. Upon switch 24 being opened(which corsponds to SCR 11 being commutated off to its blockg ornonconducting condition) the energy trapped in e filter inductance willtry to produce a coasting curnt flow in a direction such that it will bepositive at the it end of the filter inductance. This energy, which isdictly coupled across coasting diode 17, causes diode 17 to renderedconductive and to circulate a coasting curnt substantially equal to loadcurrent i;, through load 5 and coasting diode 17, thereby dischargingfilter inlctance 15. Consequently, the load voltage E and for at matterload current i will appear substantially as town in FIGURE 3(b) (2) ofthe drawings, as an esntially steady state value lower than the sourcevoltage by a factor determined by the timing of on-off switch I. Thisload voltage can be calculated from the expreson shown in FIGURE 3. Thisexpression states that the ad voltage E is equal to the time that switch24 is left osed divided by the time that switch 24 is left closed us thetime switch 24 is left open, all multiplied by the 8 power supplyvoltage E The current i supplied from the power supply to switch 24 isillustrated in FIGURE 3(b)(3) and is essentially of square wave formhaving the same period as the voltage e It should be noted that upon thenext succeeding cycle of operation when switch 24 is closed, the filterinductance 15 will again be charged in a manner such that when itdischarges upon switch 24 being opened, its potential is positive at thedot end so that the coasting rectifier 17 is again rendered conductiveand discharges the filter inductance through load 12 to provide theessentially continuous steady state load voltage E shown in FIGURE 3(b)(2).

Returning to FIGURE 1 of the drawings, it can be appreciated that thetiming of SCR11 being switched on and commutated off determines the loadvoltage E supplied across load 12 in the manner discussed in connectionwith FIGURE 3 of the drawings. In order to commutate off the SCR 11, newand improved commutation circuit means comprised by elements 16-22 hasbeen provided. The new and improved commutation circuit operates in thefollowing manner: Assuming that SCR 11 is initially in its steady stateon or conducting condition, the tap point and respective ends ofinductance 18 as well as the dot ends of capacitors 20 and 22 are eachat substantially full supply voltage. The circuit remains in thiscondition for the period of time that SCR 11 is allowed to conduct asdetermined by the time-ratio control principles described in connectionwith FIGURES 2 and 3. Thereafter, some precalculated number ofmicroseconds prior to the time that it is desired to commutate off theload current carrying SCR 11, commutating SCR 16 is turned on by theapplication of a suitable gating signal to its gate such that SCR 16conducts in a direction from the SCR end of winding 18 to power supplyterminal 14. At the same time that SCR 16 is turned on, the gatingsignal is removed from the gate of SCR 11, if it has not already beenremoved, but because SCR 11 has been conducting load current, it doesnot turn off completely instantaneously. Thus, immediately after thecommutating SCR 16 is turned on, both SCR 11 and SCR 16 are conducting,and the SCR 16 end of winding 18, which previously had been at fullsupply voltage, immediately drops to zero volts (assuming the negativesupply terminal 14 is at ground potential). In the most general case,the tap point of inductance 18 may be located at any point intermediatethe ends thereof, but for many applications it is preferably centertapped. Since the voltage across the lower half in inductance 18 rapidlyrises to the full supply voltage the SCR 11 end of winding 18 at thisinstant of time is at twice the supply voltage due to theautotransformer action of winding 18, thereby reverse biasing SCR 11.The load curren i which before the start of the commutating intervalflowed through SCR 11 and inductance 18, is switched to the cornmutatingcapacitors also immediately after the start of the commutating interval,being delayed only by circuit inductance while capacitor current reachesthe value of load current i The second condition determining the valueof the inductance of winding 18 is that it or inductors 19 and 21 besufiiciently large to limit the current inrush to capacitor 20 caused bysuch switching action. Commutating capacitors 20 and 22 are sufficientlylarge so that the voltage rise across capacitor 20, during thecommutation interval takes a sufficiently long time to charge to onehalf of the supply voltage to allow an adequate commutation time. Thecommutation interval for SCR 11 is determined by the time for capacitor20 to charge to the voltage ES/2 at which time the center tap of winding18 is also at voltage ES/2 and the SCR end thereof has been reduced toES, thereby completing the interval of reverse bias of SCR 11 so that bythis time SCR 11 should be completely commutated off. Although winning18 is ideally designed to have no leakage inductance, as a practicalmatter there will be a slight leakage inductance preseat. The leakageinductance of winding 18 and the inherent circuit inductance whichcannot be eliminated may be used in place of inductances 19 and 21.Slight oscillations of the voltage at the tap point of winding 18 bothduring and after commutation will occur at this point in the operation;however, such oscillations are rapidly damped out by the effects ofresistance in the load circuit.

After SCR 11 has been completely commutated off, capacitor 20 continuesto charge toward the voltage E and capacitor 22 to discharge toward ESCR 16 is automatically commutated off when capacitor 20 charges to avoltage greater than +E and capacitor 22 reverses in voltage due toenergy stored in the filter circuits 15, 19 and 21. At this time theexciting current in inductance 18 drops to zero and SCR 16 turns off.Diode 17 then conducts in a direction from the power supply terminal 14to the diode 17 end of winding 18. This latter conduction of di ode 17may be described as a coasting mode of operation whereby the loadcurrent is circulated within the diode 17-load circuit loop. The loadcurrent continues to circulate in the diode 17-load circuit loop due tothe energy storage within filter circuit elements 15. The advantage ofemploying a filter circuit, as shown in FIG- URES l and 3 is that loadcurrent continues to flow through load 12 even after current has ceasedto flow in the diode 17. It can be appreciated that numerous otherfilter circuits may be employed in the load circuit, however, suchfilter circuits are well known and thus will not be illustrated. If thefeedback diode 23 (shown in dotted lines in FIGURE 1) is used, thecommutation operation of the circuit of FIGURE 1 is somewhat differentfrom that described above Where diode 23- is assumed not to be presentin the circuit. With diode 23 present, during the commutating intervalof time the voltage across inductance 18 is at full supply voltage, thecenter tap point at this instant of time is one-half of the supplyvoltage. Also, since both SCR 11 (or diode 23) and SCR 16 are conductingduring this interval of time, the line current i which flows from thepositive terminal 13 passes through SCR 11, inductance 18, or capacitor20 and inductor 18 and 19 and SCR 16 to the negative power supplyterminal 14. During the commutation interval, an additional cur rent AIwill build up at a constant rate due to the impressed direct currentsupply voltage E being applied across tapped inductance 18. Inductance18, therefore, functions as a commutation interval current limitingreactor to limit DC current drawn from the power supply during thecommutation interval and is designed such that it exhibits a minimumpractical impedance to the load current while exhibiting a maximumpractical impedance to the buildup of the additional current AI duringthe commutation interval. The rate of buildup of the commutationinterval current AI is dependent upon the inductance L of the tappedwinding 18 and upon the value of the line voltage E, as set forth by theexpression di/dt E /L At the end of the commutation interval AT, thisadditional current will have the value AI=E /L AT, irrespective of thenature of the load. Accordingly, at the instant that SCR 11 iscompletely commutated off, the flux level in the core of center tappedwinding 18 has built up from its precommutation level of N L to a levelof N AI+N I and the current that had been flowing in the upper half ofwinding 18 has to be transferred to the lower half of winding 18 (or toa secondary winding as will be discussed later). At this same instant,two major conditions must be satisfied upon the current beingtransferred to the lower half of winding 18. The first of theseconditions is due to Lenzs law which requires that the flux level ininductance 18 be maintained at the value N AI-i-N I and the secondcondition requires that the commutating current flowing in inductances19 and 21 be maintained to complete the commutation interval.Accordingly, at the instant that SCR 11 completes commutation, the tappoint of inductance 18 drops from its mid-tap potential of E /Z to apotential that is determined by the requirements of Lenzs law as setforth above. For example, if the commutating interval current AI is verysmall, the voltage at the tap point will go to the negative DC potentialE thereby assuming a condition where most of the effects of commutationare largely completed. The stored energy in the inductances 19 and 21and in center tapped reactor 18 may cause a slight amount of oscillationof the voltage at the tap point, but such oscillation will die out dueto the damping effect of resistance in the load circuit. Thisoscillatory circuit is comprised by elements 20, 19, 22, 21, 15, 12, andintermittently diode 17.

If in contrast to the above-defined condition, the commutating intervalcurrent AI is appreciable or even slightly greater than the loadcurrent, the load current i;, will assume a value which will satisfy thecondition required by Lenzs law cited above. Under these circumstances,the voltage at the tap point at the time that SCR 11 is completelycommutated off will fall to a potential more negative than the value ofthe negative terminal of the direct current power supply E In the eventthat commutating interval current AI is appreciably larger than the loadcurrent, the potential at the tap point will tend to fall to a valueconsiderably below the negative terminal of the direct current powersupply E Subsequently, as the commutating interval current AI decreasesto zero, the potential of the tap point will rise toward the value ofthe negative terminal of the direct current power supply E and allow theload current i to build back up to satisfy the requirements of Lenzslaw. Upon reaching this condition, the voltage at the tap point rises tothe potential of the nega tive DC supply E and assumes a condition wherethe effects of commutation are largely completed. Again, if thecommutating interval current AI is not too great with respect to theload current, the voltage at the tap point may oscillate due to theenergy stored in the inductances 19 and 21 and reactor 18, but will bedamped out by the effects of resistance in the load circuit.

During the commutation interval, commutating capacitors 20 and 22discharge through inductances 19 and 21, respectively, since thereexists a continuously decreasing difference in potential across theseinductances during the commutation interval. The difference in potentialacross the inductances is brought about by the fact that the capacitordot ends were initially at full supply voltage +E and the voltage at thejuncture of the inductances, being at the center tap point of winding 18changed from +E to /2E at the start of the commutation interval. Thus,capacitors 20 and 22 continually discharge the inductances 19 and 21,respectively, through filter inductance 15 and load 12 during thecommutation interval. Inductances 19 and 21 effectively slow down therate of change of the potentials across capacitors 20 and 22 such thatthere is suflicient time duration to completely turn off SCR 11 beforethe dot end of the capacitors is reduced to the steady state value ofthe negative supply voltage. Since filter inductance 15 has a relativelyhigh inductance value, the load current flowing therethrough remainsrelatively constant. The total commutation time for SCR 11 is the timein which capacitors 20 and 22 discharge sufficiently such that thedischarge current flowing through inductances 19 and 21 increases toequal the load current i previouslysupplied to the load by SCR 11. Atsuch time when the full load current is supplied by the discharge ofcapacitors 20 and 22, SCR 11 no longer provides any current and,therefore, becomes nonconducting, that is, begins to be commutated off.The capacitor discharge currents flowing through inductances 19 and 21continue increasing beyond the point of equalling the load current andsuch excess current applied to inductance 19 forces diode 23 to conduct(if present). The time of such excess current flow is sufiicient topermit complete commutation of the SCR 11. After SCR 11 has becomecompletely commutated off and the total current in inductances 19 and 21is

